Asymmetric multilevel outphasing architecture for RF amplifiers

ABSTRACT

A radio frequency (RF) circuit includes a power supply configured to generate a plurality of voltages, a plurality of power amplifiers, each having an RF output port and a power supply input port, a switch network having a plurality of input ports coupled to the power supply and a plurality of switch network output ports coupled to the power supply input ports of the plurality of power amplifiers, wherein the switch network is configured to output selected ones of the plurality of voltages from the plurality of switch network output ports, at least two of the switch network output port voltages capable of being different ones of the plurality of voltages, and an RF power combiner circuit having a plurality of input ports coupled to RF output ports of the plurality of power amplifiers and an output port at which is provided an output signal of the RF circuit.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of co-pending application Ser. No.13/423,909 filed on Mar. 19, 2012 which is a continuation of applicationSer. No. 13/106,195 filed on May 12, 2011 which claims the benefit ofco-pending application Ser. No. 12/615,696 filed on Nov. 10, 2009 whichclaims priority under 35 U.S.C. §119(e) from U.S. Provisional PatentApplication No. 61/113,556, filed Nov. 11, 2008, which applications arehereby incorporated herein by reference in their entireties for allpurposes.

STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH

This invention was made with government support under contract numberFA8721-05-C-0002 awarded by the Department of the Air Force. Thegovernment has certain rights in this invention.

FIELD OF THE INVENTION

This invention relates to radio frequency (RF) circuits and moreparticularly to RF amplifier circuits.

BACKGROUND

As is known in the art, RF transmitter design is centered on a designtradeoff between the linearity of the power amplifier and itsefficiency. This tradeoff relates directly to the usefulness of theresulting device. High linearity results in a higher possible data rateand therefore compatibility with complex standards such as WirelessLocal Area Network (WLAN) and Worldwide Interoperability for MicrowaveAccess (WiMAX), and high efficiency allows for reduced cooling, energyusage and power supply requirements (e.g., in stationary applications),and longer use or smaller battery size (e.g., in cell phone and portableapplications). The general perception that the tradeoff betweenlinearity and efficiency is fundamental tends to produce designs thatcompromise between these two design goals. The resulting systems may beeither linear or efficient, or are designed specifically for a singlecommunications standard and therefore have limited flexibility of use.Meanwhile, consumer demand for both greater transmission rates andsmaller devices continues to drive the need for an architecture that iscapable of both linearity and efficiency.

As is also known in the art, communications standards that support highdata rates such as WLAN/WiMAX employ variable-envelope modulation, andso linear amplification is required. One conventional approach is to usean inefficient but highly linear power amplifier. However, there are twomain types of transmitter architectures that enable the use of moreefficient but non-linear switching mode power amplifiers: (1) polar, and(2) outphasing, or linear amplification of nonlinear components (LINC).

Conventional polar architectures divide a signal to be amplified intoamplitude and phase components. The phase component is used as the inputto a non-linear, high-efficiency switching power amplifier, while theamplitude component drives the power supply of the power amplifier tocreate a varying-envelope signal. While this improves the poweramplifier efficiency, it also requires the use of an efficientwide-output range, high-bandwidth power converter. Because converterefficiency degrades dramatically as bandwidth increases, it is verydifficult to achieve high efficiency for high data-rate communicationstandards. This is exacerbated by the 5-10× bandwidth expansion thatoccurs during the conversion from Cartesian to polar coordinates. Thus,this conventional approach is only practical for low-bandwidth systems.

Outphasing, and specifically conventional LINC architectures, is basedon the fact that an arbitrary input signal that can be divided into twoconstant-amplitude, phase-modulated signals that can each benon-linearly amplified and then passively recombined as a vector sum.This vector sum produces an output signal that is a linearly amplifiedversion of the input. The LINC strategy eliminates the high-bandwidthpower converter of the polar architecture, using outphasing to realizeamplitude variation. However, the efficiency of the power combining ishigh only over a small range of output powers. To avoid signaldistortion and preserve amplifier efficiency, an isolating combiner isoften used. Conventional isolating combiners achieve 100% efficiencyonly at maximum output power. When the inputs are outphased to vary theamplitude, power is wasted as heat in the isolation resistor. The resultis an overall efficiency that is inversely proportional to thepeak-to-peak average power ratio (PAPR), limiting the benefits of thisconventional approach in high data-rate communication standards such asWiMAX, in which the PAPR is high.

One of the major drawbacks of the LINC architecture is the power wastedin the power combiner. However, a combiner must be used to isolateoutphased power amplifiers and provide a fixed impedance load to thepower amplifiers in order to avoid signal distortion and preserveswitching amplifier efficiency. But power is wasted as heat in thecombiner resistor when the inputs are outphased to vary the amplitude.Since the power delivered to the combiner by power amplifiers isconstant, the efficiency of the LINC system is directly proportional tothe output power sent to a load. The time-averaged efficiency istherefore inversely proportional to the peak-to-average power ratio(PAPR). Unfortunately, high-level modulation schemes such as 64-QAM andOFDM tend to have high PAPR, leading to low average efficiency when theLINC system is used.

To alleviate the problem of wasted energy during outphasing, sometimesnon-isolating combiners are used. The Chireix combiner is a prominentexample which uses compensating reactive elements to enhance thepower-combining efficiency. However, the Chireix combiner can only betuned for a very small range of outphase angles. With outphase anglesoutside the tuned range, the load impedance presented to the poweramplifiers deviates too far from the nominal value and the isolationbetween the power amplifier outputs becomes poor. The result issignificant distortion and degraded amplification efficiency.

One proposed power recycling technique described in Zhang X., at al.“Analysis of power recycling techniques for RF and microwave outphasingpower amplifiers,” IEEE Trans. Circuit Syst. II, vol. 49, no. 5, May2002, pp. 312-320, attempts to enhance the power efficiency of the LINCarchitecture without giving up the simplicity of an isolating combiner.The isolation resistor is replaced with an RF-dc converter to recoverthe wasted power back to the power supply. While this approach has beenshown to result in a significant increase in the overall efficiency, itsuffers from excessive impedance variation at the isolation port andtherefore incomplete isolation between power amplifiers. This can leadto excessive signal distortion and lower efficiency or even completebreakdown in the power amplifiers, particularly in those sensitive toload impedance, such as many switched-mode power amplifiers. Anadditional isolator can be added between the isolation port and theRF-dc converter to reduce this effect, but at the cost of addedcomplexity and loss.

SUMMARY

In general overview, the circuits, concepts, and techniques describedherein provide an asymmetric multilevel outphasing (AMO) transmitterarchitecture which includes a switch network capable of supplyingdiscrete voltages to power amplifiers. The power amplifiers are powercombined to provide an output signal over a wide power range. An AMOmodulation technique is used to minimize amplifier outphasing angles toachieve higher efficiency and linearity in a transmission architecture.The result is a highly efficiency architecture that is compatible with awide range of communication standards and applications. For example, theinventive concepts, circuits, and techniques described herein may beused to provide highly efficient military and/or commercial transmittersfor use in handheld units, laptop wireless modems, and base stations.The inventive concepts, circuits and techniques may also be used toprovide high-efficiency RF power amplification for medical applications(such as RF amplifiers for Magnetic Resonance Imaging) and industrialand commercial applications (such as plasma generation, heating,coating, and sintering),

In one aspect, a radio frequency (RF) circuit includes a power supplyconfigured to generate a plurality of voltages, a plurality of poweramplifiers, each having an RF output port and a power supply input port,a switch network having a plurality of input ports coupled to the powersupply and a plurality of switch network output ports coupled to thepower supply input ports of the plurality of power amplifiers, whereinthe switch network is configured to output selected ones of theplurality of voltages from the plurality of switch network output ports,at least two of the switch network output port voltages capable of beingdifferent ones of the plurality of voltages, and an RF power combinercircuit having a plurality of input ports coupled to RF output ports ofthe plurality of power amplifiers and an output port at which isprovided an output signal of the RF circuit.

In further embodiments, the RF circuit includes one or more of thefollowing features: each of the power amplifiers has an RF input portconfigured to receive a phase-adjusted signal and the switch network isconfigured to receive a plurality of control signals, wherein thephase-adjusted signals and the switch network control signals are usedto control the output signal of the RF circuit; a control systemconfigured to provide the phase-adjusted signals over a plurality offirst output ports coupled to the RF input ports of the plurality ofpower amplifiers and the plurality of control signals over a pluralityof second output ports coupled to the switch network; the control systemis further configured to decrease a difference between a total of thepower output from the plurality of power amplifiers and a power outputfrom the RF circuit; the RF power combiner circuit includes an isolatingcombiner; the power combiner circuit further includes a resistancecompression network and a rectification circuit coupled to theresistance compression network; a plurality of LC filters configured tocouple the switch network to respective ones of the plurality of poweramplifiers, and; at least one of the power amplifiers is width-switched.

In another aspect, the concepts, circuits, and techniques describedherein are directed to an outphasing energy recovery amplifier (OPERA)architecture that substantially reduces the impedance variation at anisolation port of a combiner through the use of a resistance compressionnetwork (RCN). The RCN improves the matching and isolation betweenoutphased amplifiers, helping to maintain high linearity as well as highefficiency in switching-mode amplifiers.

The OPERA architecture includes circuitry to recover power that wouldotherwise be wasted in an isolation resistor back to a power supply. Insome embodiments, the isolation resistor is replaced with an RF-dcconverter. The equivalent input impedance of the rectifier varies withinput power which can reduce the isolation between the power amplifiersand can lower power amplification efficiency (and in some instances,cause complete malfunction) and increase unwanted signal distortion atthe output. To mitigate these unwanted effects, a RCN is included toreduce the rectifier impedance variation. In some embodiments, animpedance transformation stage is placed between the RCN and the powercombiner's isolation port to match the resistance-compressed rectifierimpedance to the impedance required by the power combiner.

In another aspect, a radio frequency (RF) circuit includes a pluralityof power amplifiers, each having an RF output port, and an RF powercombiner circuit having a plurality of input ports coupled to respectiveones of the RF output ports of the plurality of power amplifiers, andincluding a resistance compression network, a rectification circuitcoupled to the resistance compression network, and an output port atwhich is provided an output signal of the RF circuit.

In further embodiments, the RF circuit includes one or more of thefollowing features: the RF power combiner circuit provides isolationbetween the plurality of RF power combiner circuit input ports; each ofthe power amplifiers has a power supply input port and further includinga power supply providing voltages at a plurality of power supply outputports, wherein at least two of the power supply output ports providedifferent voltages, and a switching circuit to selectively couple eachpower amplifier power supply input port to at least one of the powersupply output ports, and; each of the power amplifiers has an RF inputport configured to receive a phase adjusted signal and the switchingcircuit is configured to receive at least one control signal, whereinthe phase adjusted signals and the at least one switching circuitcontrol signal are used to control the output signal of the RF circuit.

In a further aspect, a radio frequency (RF) transmission method includesproviding a plurality of voltages, outputting selected ones of theplurality of voltages to a plurality of power amplifiers, at least twoof the outputted voltages capable of being different ones of theplurality of voltages, and providing an RF power combiner circuit havinga plurality of input ports coupled to the plurality of power amplifiers,and an output port at which is provided an output signal of the RFcircuit.

In further embodiments, the method includes one or more of the followingfeatures: providing a power supply to generate the plurality ofvoltages, providing the plurality of voltages to a plurality of inputports of a switch network, and in the switch network, providing theselected ones of the plurality of voltages to RF input ports of theplurality of power amplifiers; each of the power amplifiers has an RFinput port, further including controlling the output signal of the RFcircuit by adjusting the phases of the signals received at the RF inputports and dynamically selecting the ones of the plurality of voltagesoutputted to each of the power amplifiers; decreasing a differencebetween a sum of the powers outputted by the power amplifiers and an RFpower outputted at the output port of the RF circuit; said decreasingincludes minimizing the difference between the sum of the powersoutputted by the power amplifiers and the RF power outputted at theoutput port of the RF circuit; gating on a variable number oftransistors in at least one of the power amplifiers; providing isolationbetween the plurality of input ports, and; processing at least a portionof the RF power output from the power amplifiers using at least oneresistance compression network and at least one rectification circuitcoupled to the at least one resistance compression network, wherein theprocessed RF power includes recovered RF power from the RF powercombiner circuit.

BRIEF DESCRIPTION OF THE DRAWINGS

The foregoing features of concepts, circuits, and techniques describedherein may be more fully understood from the following description ofthe drawings in which:

FIG. 1 is a block diagram of an embodiment of an asymmetric multileveloutphasing (AMO) circuit;

FIG. 1A is a circuit diagram of an embodiment of a switch network as maybe used in the AMO circuit of FIG. 1;

FIG. 1B is a circuit diagram of another embodiment of a switch networkas may be used in the AMO circuit of FIG. 1;

FIG. 1C is an exemplary embodiment of a power supply as may be used forsupplying voltages to the AMO circuit of FIG. 1;

FIG. 2 is a schematic circuit diagram of an M-way power amplifier/N-wayvoltage level AMO circuit embodiment;

FIG. 3 is a block diagram of 4-way power amplifier/4-way voltage levelAMO circuit embodiment including a 4-way matched combiner;

FIG. 4 is a graph showing power efficiency curves of AMO circuitembodiments having 2-way voltage levels;

FIG. 5 is another graph showing power efficiency curves of further AMOcircuit embodiments having 4-way voltage levels;

FIG. 6 is a circuit diagram of a width-switched power amplifierembodiment;

FIG. 7A is a block diagram of a two power amplifier/four voltage levelcircuit embodiment including a control system;

FIG. 78 is polar coordinate graphical representation of coordinates (I,Q) of a baseband signal;

FIG. 8 is a block diagram of an embodiment of an outphasing energyrecovery amplifier;

FIG. 9 is a block diagram of an embodiment of a resistance compressionnetwork embodiment used in the amplifier embodiment of FIG. 8; and

FIG. 10 is a flow diagram of an asymmetric multilevel outphasingtransmission method.

DETAILED DESCRIPTION

Referring now to FIG. 1, in one aspect a radio frequency (RF) circuit100 includes a power supply 110 configured to generate a plurality ofvoltages V₁, V₂, V₃-V_(N) (generally designated by reference numeral115), a plurality of power amplifiers 120A, 120B-120N (generallydesignated by reference numeral 120), each having an RF output port122A, 122B-122N (generally designated by reference numeral 122) and apower supply input port 124A, 1248-124N (generally designated byreference numeral 124). The RF circuit 100 includes a switch network 130having a plurality of input ports (generally designated by referencenumeral 132) coupled to the power supply 110 and a plurality of switchnetwork output ports (generally designated by reference numeral 134)coupled to the power supply input ports 124 of the plurality of poweramplifiers 120.

In the RF circuit embodiment of FIG. 1, switch network 130 includesswitch circuits 130A, 130B-130N each of which is coupled to respectivepower amplifiers 120A, 120B-120N. Each of the switch circuits 130A,130B-130N includes a number of switches (generally designated byreference numeral 136) selectively coupled to respective input voltagesV₁, V₂, V₃-V_(N) to output selected ones of the voltages 115. Forexample, each of the switch circuits 130A, 130B-130N includes fourswitches 136 to select one of the four input voltages V₁, V₂, V₃-V_(N).It should be noted that although four input voltages (and fourrespective switches) are shown, one of ordinary skill in the art willreadily appreciate that any number of input voltages may be used, forexample, two, three, five, ten, 100, 1000, etc., and switch network 130may be provided including an appropriate number of switch network inputports and switch network output ports.

The switch network 130 (which in some embodiments may be referred to asa switching circuit) is configured to output selected ones 116A,116B-116N (generally designated by reference numeral 116) of theplurality of voltages 115 at the plurality of switch network outputports 134. At least two (i.e., two, three, five, ten, 100, 1000, etc.)of the switch network output port voltages 134 are capable of beingdifferent ones of the plurality of voltages 115. As by way of anon-limiting example shown in the RF circuit embodiment of FIG. 1, threeof the selected voltages 116A, 116B, and 116N are different voltages,namely respective input voltages V₁, V₂, and V_(N).

It should be noted that the selected voltages 116 need not be different.For example, a single voltage (e.g., V₁) may be selected for output atthe switch network output ports 134. In other words, even though theswitch network 130 is capable of outputting different ones of the inputvoltages 115, the same input voltage may be selected for output at theswitch network output ports 134.

The RF circuit 100 further includes an RF power combiner circuit 140having a plurality of input ports 142A, 142B-142N (generally designatedby reference numeral 142) coupled to RF output ports 122 of theplurality of power amplifiers 120, and an output port 144 at which isprovided an output signal S_(out) of the RF circuit 100. In a furtherembodiment, the RF power combiner 140 is an isolating combiner.

In another embodiment, the RF circuit 100 includes a plurality oflow-pass filters coupled between the switch network 130 and the poweramplifiers 120. The low-pass filters can provide pulse shaping to reduceor in some cases minimize and/or even eliminate undesirable highfrequency content that may be introduced into a signal primarily causedby rapid changes in the switched supply voltages 115. In someembodiments, these low-pass filters are nominally low-order LC filterswith low loss, but there are many different ways that a low-pass filtercan be implemented. For example, another possibility is that theparasitic capacitances and inductances, always present in any physicalcircuit, provide enough filtering that an explicit low-pass filter isnot required. A further possibility is that the energy storage of the RFpower amplifiers 120 themselves (such as owing to the use of RF inputchokes or inductors) may provide enough filtering that an explicitlow-pass filter is not required.

In one or more embodiments, the RF circuit 100 may be referred to as anasymmetric multilevel outphasing (AMO) architecture for multi-standardtransmitters. The AMO architecture can be generalized to include two ormore power amplifiers, as may be similar to power amplifiers 120described in conjunction with FIG. 1. When combined, such two or morepower amplifiers are herein referred to as an “M-way” power amplifiers.An output of M-way power amplifiers may be described as a vector sum ofM different power amplifier outputs, each of which can have two or moredifferent supply voltages, as may be similar to input voltages 115described in conjunction with FIG. 1. Such two or more supply voltages,when combined, are herein referred to as “N-way” supply voltages.Furthermore, each of the M power amplifiers may have an arbitrary phase.

In further embodiments, the RF circuit 100 includes a control system 150further described herein below.

It will be appreciated by one of ordinary skill in the art that the RFcircuit 100 is not limited to switch circuits 130A, 130B-130N forselecting input voltages 115. As by way of non-limiting examples, amultiplexor circuit may be used to select the input voltages 115 foroutput to the power amplifiers 120.

Referring now to FIG. 1A, a further embodiment of a switch network 130′includes one or more switch circuits 130A′, each of which is coupled oneof the power amplifiers 120 (shown in FIG. 1).

Referring now to FIG. 1B, another embodiment of a switch network 130″includes one or more switch circuits 130A″, each of which is coupled toone of the power amplifiers 120. It will be understood by one ofordinary skill in the art that some embodiments of a switch network mayinclude combinations of switch circuits (e.g., combinations of switchcircuits 130A, 130A′, and/or 130A″).

Referring again to FIG. 1, it should be noted that power supply 110 isnot limited to any particular type of power supply and includes most anypower supply capable of generating the plurality of voltages 115.Referring now to FIG. 1C, a non-limiting example of a switched-capacitorpower supply 110′ is shown including a voltage supply 111, switches (anexample of which is designated by reference numeral 112, and switchedcapacitors (an example of which is designated by reference numeral 113)to provide voltages 115′. It should be noted that although four voltagesare shown, the power supply 110′ may generate any number of neededand/or desired voltages.

Referring now to FIG. 2, a schematic circuit diagram of a discretesupply-modulated power amplifier circuit 220 includes a power amplifier220A coupled through switches 230 to voltage supplies V_(sup1),V_(sup2)−V_(supN). The power amplifier 220A receives an arbitrary phasesignal φ_(in,x).

The discrete supply-modulated power amplifier circuit 220 may berepresented as an equivalent circuit layout 260, which includes avoltage supply 262, resistor 264, and output voltage V_(x) 266. Aschematic of an M-way AMO power amplifier circuit 270 includes M circuitlayouts 260 (an example of which is designated by reference numeral220′) coupled in parallel to a matched, lossy, M-way combiner 280providing output voltage V_(out).

Referring now to FIG. 3, an example of an M-way, N-way circuit 300 isshown in which M=4 and N=4. A 4-way matched combiner 380 combinesoutputs 322 of each of the power amplifiers 320. The 4-way matchedcombiner 380 is realized as a corporate array (or binary tree) of 2-wayWilkinson combiners.

It will be appreciated by one of ordinary skill in the art that othertypes of combiners may be used. As by way of non-limiting examples, acombiner may include a binary or “corporate tree” of 2-way combiners, anM-way Wilkinson combiner, and/or a M-way inter-phase transformer withisolation resistors.

An M-way AMO circuit of the type described herein can be advantageous athigh frequencies and power levels. For example, using two or moreoutphased power amplifiers in an AMO circuit can increase the number ofefficiency peaks in power output performance for a given number ofsupply voltage levels. The efficiency for a given supply voltagecombination using a matched isolating M-way combiner can be calculatedas follows:

$\begin{matrix}{\eta = \frac{P_{out}}{\sum\limits_{k = 1}^{M}P_{k}}} \\{= \frac{{V_{out}}^{2}}{\sum\limits_{k = 1}^{M}{V_{k}}^{2}}}\end{matrix}$

Here, P_(k) is the output power of the k^(th) power amplifier, V_(k) isthe output voltage of the k^(th) power amplifier, P_(out) is the outputpower, and V_(out) is the output voltage. This assumes 100% efficientpower amplifiers and no combiner insertion loss. Note that if asymmetric dissipative isolating combiner is used, 100% efficiency canonly be obtained when all the voltages being combined have the sameamplitude. Therefore, there will be exactly N points of 100% efficiencyin power output performance. When the voltages being combined havedifferent amplitudes, there is loss in the combiner's isolationresistors.

Referring now to FIG. 4, a graph 400 has a horizontal axis denotingnormalized output power in units of decibels (dB) and a vertical axis inpercentage of power efficiency. In graph 400, theoretical powerefficiency curves 402, 404, 406 are shown for respective M-way AMOcircuits in which M=2, 3, 4, respectively, and in which N=2 voltagesupply levels. A theoretical power efficiency curve 410 is also shownfor a conventional linear amplification using non-linear components(LINC) circuit. The power efficiency curve for a given value of M (i.e.,M=2, M=3, etc.) may be generated by first computing the efficiency vs.output power for each possible voltage combination, setting theefficiency to 0 if a given output power is unachievable for a givenvoltage combination, and taking the maximum efficiency over thedifferent possible voltage combinations. Supply voltages have beenselected such that two 100% efficiency points are separated by 6 dB. Ascan be readily seen in FIG. 4, a number of power efficiency peaks (anexample of which is denoted by reference numeral 411) increases as Mincreases.

Referring now to FIG. 5, a graph 500 has a horizontal axis denotingnormalized output power in arbitrary units of decibels (dB) and avertical axis in percentage of power efficiency. In graph 500,theoretical efficiency curves 502, 504, 506 are shown for respectiveM-way AMO circuits in which M=2, 3, 4 and in which N=4 voltage supplylevels. Also shown is a theoretical power efficiency curve 510 for aconventional LINC circuit. Supply voltages have been selected such thatfour 100% efficiency points are separated by 3 dB. As can be readilyseen in FIG. 5, a number of power efficiency peaks (an example of whichis denoted by reference numeral 511) increases as M increases.

For a given output voltage vector V_(out)=A·exp(j·θ) and a givencombination of power amplifier supply voltages, the phases for each ofthe power amplifiers can be computed as described herein below.

An output voltage may be defined as a vector sum of the M voltagevectors from each power amplifier as follows:{right arrow over (V)} _(out) ={right arrow over (V)} ₁ +{right arrowover (V)} ₂ + . . . +{right arrow over (V)} _(M) =A∠θ

The output voltage vector can be separated into real and imaginarycomponents as follows:Re({right arrow over (V)} _(out))=|V ₁|cos φ₁ +|V ₂| cos φ₂ + . . . +|V_(M)| cos φ=A cos θIm({right arrow over (V)} _(out))=|V ₁|sin φ₁ +|V ₂| sin φ₂ + . . . +|V_(M)| sin φ_(M) =A sin θ

These two equations yield M unknowns, which are the phases of the Mpower amplifiers. There are multiple possible solutions for M phasesand, in some cases, no solution exists for a given amplitude A and agiven set of voltage levels V_(k). For purposes of illustration, theoutphasing angles and voltage supply levels are calculated in such wayas to minimize energy loss. Described here is method for the case ofM=2. However, it should be understood that the method can be generalizedto handle cases for which M>2.

In order to achieve an output vector with amplitude A, let the outputamplitude of one power amplifier be A₁ chosen from a discrete set ofpossible values V_(k), and that of the other be A₂, also chosen from thesame set of discrete possible values. For each possible value of A₁ andA₂, the efficiency of the power combining operation can be calculatedusing the formula:

$\eta_{c} = \frac{A^{2}}{A_{1}^{2} + A_{2}^{2}}$

All combinations of A₁ and A₂ for which this formula evaluates to avalue exceeding 1 are impossible choices for realizing the outputamplitude A. The values of A₁ and A₂ for which t is maximized (withoutexceeding 1) are the most efficient choices. That is, they result in theminimum outphasing angle and the minimum amount of wasted energy. Oncethe values A₁ and A₂ are chosen, the proper phases for the two poweramplifiers are given by the following equations:

${\varphi_{1}(t)} = {{\theta(t)} + {\cos^{- 1}\frac{\left( {{V_{1}(t)}^{2} + {2{A(t)}^{2}} - {V_{2}(t)}^{2}} \right)}{2\sqrt{2}{V_{1}(t)}{A(t)}}}}$${\varphi_{2}(t)} = {{\theta(t)} - {\cos^{- 1}{\frac{\left( {{V_{2}(t)}^{2} + {2{A(t)}^{2}} - {V_{1}(t)}^{2}} \right)}{2\sqrt{2}{V_{2}(t)}{A(t)}}.}}}$

In an AMO power amplifier circuit, as may be similar to RF circuitembodiment 100 described in conjunction with FIG. 1, output power andcircuit conduction current levels change with the supply voltagesdelivered to the power amplifiers. The circuit conduction losses andswitching losses decrease because the power supply input is switched toconsecutively lower voltages as power is reduced. Gate drive power,however, does not experience similar reductions with output power, whichcan negatively impact efficiency at low output power levels.

Referring now to FIG. 6, in some embodiments, an AMO power amplifiercircuit includes a width-switching device 600 in a switching poweramplifier 620, such as a class E switching amplifier. Such a device canparallel the output (drain-source) ports of multiple transistors 611 anddrive transistor gates 612 with separately controlled gate drives 614.At low power levels, some of the gate drives 614 can be disabled (orotherwise driven to leave transistors 611 off) to save gating power. Insuch a way, gate drive loss reduction may be traded off for increase inon-state conduction, which can allow optimization of the number oftransistor elements gated as a function of power level. Moreparticularly, a number and relative size of width-switching devices 600can be provided and driven separately at an input source (and operatedin parallel at transistor outputs) so as to provide good efficiency overa desired power range.

In an exemplary operation of width-switching device 600, when V_(in) isrelatively large (for example, selected as a large input voltage forhigh power output), a first gate drive (i.e. gate drive 1) and a secondgate drive (i.e., gate drive 2) provide AC gate-drive switching signalsto transistors 611. Alternatively, when V_(in) is relatively small (forexample, selected as a small input voltage for lower power output) oneof the gate drive switching signals is modified to hold the gate driveoutput low to deactivate one of the transistors while another one oftransistor is gated on and off.

In a further embodiment, first and second gate drives providesubstantially similar gating patterns.

In another embodiment, at least one of the gate drives is a plurality ofcoupled amplifiers.

In a further embodiment, more than two width-switching devices could besized equally in a geometric sizing arrangement (e.g., widths A, 2A, 4A,etc.) or other sizing strategy. In still further embodiments, devicesare matched to realize an optimum lowest loss for different poweramplifier input voltages of the AMO circuit. This can enable highefficiency at each power supply level in the AMO circuit.

Referring again to FIG. 1, in a further RF circuit embodiment each ofthe power amplifiers 120A, 120B-120N has an RF input port 126A,126B-126N (generally designated by reference numeral 126) configured toreceive respective phase-adjusted signals φ₁, φ₂−φ_(N) (generallydenoted by reference numeral 135). Furthermore, the switch network 130is configured to receive a plurality of control signals V_(C1),V_(C2)−V_(CN) (generally designated by reference numerals 125). As willbe described herein below, the phase-adjusted signals 135 and controlsignals 125 control the output signal 144 of the RF circuit.

In still a further embodiment, a control system 150, which receives asinput an amplitude A and a phase φ, is configured to provide thephase-adjusted signals 135 over a plurality of first output ports 154coupled the RF input ports 126 of the power amplifiers 120 and thecontrol signals 125 over a plurality of second output ports 152 coupledto the switch network 130.

Referring now to FIG. 7A, in a further embodiment an AMO circuit 700includes a control system 750 and an RF circuit 701. The control system750 includes a predistorter 760, an AMO modulator 770, and a digitalradio frequency power converter (DRFPC) 780 for modulating a basebandsignal comprising amplitude A and phase φ components. The RF circuit701, which may be a further embodiment of the RF circuit embodiment 100described in conjunction with FIG. 1, includes a first switch 730A and asecond switch 730B for selecting voltage levels 715 supplied torespective first power amplifier 720A and second power amplifier 720B.The voltage levels 715 are received from a power supply (not shown) asmay be similar to power supply 110 described in conjunction with FIG. 1.In a further embodiment, the AMO modulator 770 drives a fast switchingnetwork and switching mode amplifiers, which may include, but are notlimited to, class-E, class-F, class-φ, and/or class E/F poweramplifiers. An RF power combiner 740 combines the outputs of first andsecond power amplifiers 720A, 720B, while providing isolation betweenits input ports.

The predistorter 760 linearizes the combined non-linearity from theDRFPC 780, switches 730A, 730B, and power amplifiers 720A, 720B. A polarlookup table 762 is used to store lookup values for amplitude A andphase φ components as will be described herein below. The AMO modulator770 determines a combination of two power voltages 715 supplied to thepower amplifiers 720A, 720B based on a peak amplitude within a timeinterval, which in a further control system embodiment is determined ina interval peak detector. The AMO modulator 770 decomposes apredistorted amplitude and phase received from the predistorter 760 intoa pair of amplitude values (A₁, A₂) and a pair of phase values (φ₁, φ₂)using a first-order approximation of equations 3A and 3B describedherein below. In a further embodiment, the AMO modulator includes a timealigner 772 to maintain any time delay mismatch between amplitude paths773 and phase paths 774 to within the margin required by a particularapplication.

The DRFPC 780 performs phase modulation by embedding phase componentsφ₁, φ₂ of the AMO modulator output into an RF carrier signal. The DRFPC780 includes an array of current steering switches and can bring asignificant transmitter power efficiency boost particularly for lowoutput power levels for two reasons. First, the analog matchingrequirement in the current steering switches is relaxed because thestatic phase errors in the DRFPC output, which result from analogmismatch, can be corrected by the predistorter 760. Second, the DRFPC780 does not need baseband active filters for DAC output shaping.

Referring now to FIG. 7B, an exemplary operation of an asymmetricmultilevel outphasing (AMO) modulation technique to determine controlvoltages and phase components will now be described. A graph 790 is apolar representation in coordinates (I, Q) of a baseband signal. Halfcircles (an example of which is designated by reference numeral 792)correspond to discrete amplitude values. The graph 790 includes acomplex vector 793 at a phase-amplitude baseband constellation point793A.

The AMO modulation technique decomposes the complex vector 793 into afirst vector 795 and a second vector 797. The first and second vectors795, 797 are a baseband representation of outputs of power amplifiers,as may be similar to power amplifiers 720A and 720B of the RF circuitembodiment 701 described in conjunction with FIG. 7A. An outphasingangle θ is defined between the first and second vectors 795, 797.

Mathematically, AMO modulation technique can be defined with the polarrepresentation of the baseband signal, according to the followingequation:C(t)=r _(i)(t)+jr _(q)(t)=A(t)e ^(jθ(t))  (1)

Here, C represents a baseband signal over time t, and r_(i) and r_(j)are respective real and imaginary coordinates of baseband signal C. Inequation (1), A represents amplitude and θ represents the angle.

C(t) can be linearized by predistorting power amplifier output using apolar lookup table (as may be similar to polar lookup table 762described in conjunction with FIG. 7A) using the following equation:P(t)=A _(p)(t)e ^(jθp(t))  (2)

Here, θ_(p) is the lookup table value. In an RF circuit including afirst and a second power amplifier (as may be similar to RF circuit 701described in conjunction with FIG. 7A), P(t) can be decomposed into twoparts using the following equation:P(t)=W(V ₁(t)e ^(jφ1(t)) ,V ₂(t)e ^(jφ2(t))  (3A)

Here, V₁ represents a first voltage level output at time t from thefirst power amplifier and proportional to the input power supply voltageinto the first power amplifier and V₂ represents a second voltage leveloutput at time t from the second power amplifier and proportional to theinput power supply voltage into the second power amplifier. W representsWilkinson power combining. In this way, voltage levels (i.e., firstvoltage level and second voltage level) can be dynamically selected overtime and/or at various times during operation of the AMO circuit.Advantageously, the AMO circuit is able to adjust to dynamicpower-efficiency needs of an application.

A first phase component φ₁ representing a first phase input to the firstpower amplifier and a second phase component φ₂ representing a secondphase input to a second power amplifier can be calculated as follows:

$\begin{matrix}{{{\Phi_{1}(t)} = {{\theta_{p}(t)} + {\cos^{- 1}\frac{\left( {{V_{1}(t)}^{2} + {2{A_{p}(t)}^{2}} - {V_{2}(t)}^{2}} \right)}{2\sqrt{2}{V_{1}(t)}{A_{p}(t)}}}}},{{\Phi_{2}(t)} = {{\theta_{p}(t)} - {\cos^{- 1}\frac{\left( {{V_{2}(t)}^{2} + {2{A_{p}(t)}^{2}} - {V_{1}(t)}^{2}} \right)}{2\sqrt{2}{V_{2}(t)}{A_{p}(t)}}}}}} & \left( {3B} \right)\end{matrix}$

The AMO modulation technique can be used to optimize efficiency of an RFcircuit (as may be similar to RF circuit embodiment 100 described inconjunction with FIG. 1) by minimizing power loss in a power combiner(as may be similar to RF power combiner 140 described in conjunctionwith FIG. 1). An optimal value of each level r_(k) can be determined, inwhich levels r_(k) are the maximum output amplitudes A for each supplyvoltage levels when a power supply drives power amplifiers (as may besimilar to power supply 110 and power amplifiers 120A and 120B describedin conjunction with FIG. 1). The Wilkinson combiner efficiency at agiven output amplitude A driven by two power amplifiers with differentsupply voltages can be represented according to the following equation:

$\begin{matrix}{{\eta_{c}\left( {A,r_{k},r_{j}} \right)} = \frac{A^{2}}{r_{k}^{2} + r_{j}^{2}}} & (4)\end{matrix}$

Equation (4) simplifies to a standard Wilkinson efficiency whenr_(k)=r_(j). The total average efficiency can be computed if theamplitude power distribution function (PDF) p(A) of the signal is known.For example, total average efficiency can be computed by dividing thePDF into several regions separated by the r_(k) (and r_(k)combinations), integrating the PDF curve to find the efficiency in eachregion, and summing the result. For N different supply voltages, therewill be

$\begin{pmatrix}N \\2\end{pmatrix}\quad$combination of supply voltages given two power amplifiers. However, thepower combiner efficiency decreases as the difference between twovoltage levels increases. Also, the efficiency improvement may berelatively small when the difference between the two voltages isrelatively large. Therefore, the supply voltage combinations can berestricted to adjacent voltage supply levels (i.e., r_(k) and r_(k+1)).Using this restriction together with the known PDF of the transmittedsignal, the optimum combination of supply voltages can be determined byexhaustive search.

Although AMO modulation has been described using Wilkinson powercombining, one of ordinary skill in the art will readily appreciate thatother power combining techniques may be used. Furthermore, although AMOmodulation has been described with reference to two power amplifiers,such is not intended as limiting and one of ordinary skill in the artwill readily appreciate that more than two power amplifiers may be used.

Referring now to FIG. 8, in another aspect an RF circuit 800 includes aplurality of power amplifiers 820, each having an RF output port 822 andan RF power combiner circuit 840 having a plurality of input ports 842coupled to respective ones of the RF output ports 822 of the pluralityof power amplifiers 820. The RF power combiner 840 includes a resistancecompression network (RCN) 860, a rectification circuit 865 coupled tothe resistance compression network 860, and an output port 844 at whichis provided an output signal S_(out) of the RF circuit 800. The RFcircuit output port 844, in some embodiments, is coupled to a load 811,such as an antenna.

In a further embodiment, an impedance transformation stage 868 iscoupled to an isolation port 848 of the power combiner 840 and the RCN860. The impedance transformation stage 868 matches a RCN impedance toan impedance required by the power combiner 840.

The RF circuit embodiment 800 of FIG. 8 includes a first power amplifier820A and a second power amplifier 820B. In some embodiments, the firstpower amplifier 820A receives a first signal S₁(t) output from modulator821A and the second power amplifier 820B which receives a second signalS₂(t) output from modulator 821B. In the same or different embodiment, asource signal S(t) may be fed through a sinusoidal signal source (SCS)to provide signals S₁(t) and S₂(t). A voltage supply 823 provides powerto each of the power amplifiers 820A, 820B and recovers power from RFpower combiner 840 as will be described herein below.

An exemplary operation of the RF circuit embodiment 800 will now bedescribed. Because the power combiner 840 requires a fixed resistance atthe isolation port 848 to ensure matching and isolation between thefirst and second outphased power amplifiers 820A, 820B, the RF-dcconverter which recovers the wasted power should provide a constantresistive impedance at its input. A purely resistive input impedance canbe achieved with a variety of rectifier structures, a non-limitingexample of which includes an ideal half bridge rectifier driven by asinusoidal current source of amplitude I_(in) and frequency ω_(s), andhaving a constant output voltage V_(dc). A voltage at the inputterminals of the rectifier V_(x)(t) will be a square wave having afundamental component of amplitude V_(x1)=(2V_(dc)/π) in phase with aninput current i_(in)(t). The electrical behavior at the fundamentalfrequency ω_(s) (neglecting harmonics) can be modeled as a resistor ofvalue R_(eq)=(2/π) (V_(dc)/I_(in)). One of ordinary skill in the artwill readily appreciate that there are many other types of rectifiertopologies that can achieve the above-mentioned behavior.

Driving a rectifier (such as the above-described ideal half bridgerectifier) with a tuned network suppresses the harmonic content inherentin rectifier operation and results in a resistive impedancecharacteristic at a desired frequency. This equivalent resistance can berepresented by the following equation:

$\begin{matrix}{R_{rect} = {k_{rect}\frac{V_{dc}}{I_{1}}}} & (5)\end{matrix}$where k_(rect) depends on the specific rectifier structure and |I₁| isthe fundamental component of the drive current. Ignoring harmonics, thepower delivered to the rectifier is P_(in)=½I_(in) ² R_(rect). Therectifier impedance can be written as follows:

$\begin{matrix}{R_{rect} = \frac{\left( {k_{rect}V_{dc}} \right)^{2}}{2P_{in}}} & (6)\end{matrix}$

Equation (6) shows that the rectifier input impedance is inverselyproportional to input power. The equivalent input impedance of therectifier varies with input power which can reduce the isolation betweenthe power amplifiers and can lower power amplification efficiency (andin some instances, cause complete malfunction) and increase unwantedsignal distortion at the output.

To mitigate these unwanted effects, an RCN 860 is included to reduce therectifier impedance variation. The RCN 860 can be combined with anappropriate set of rectifiers 865 to yield an RF-dc converter withnarrow-range resistive input characteristics.

Although operation of the outphasing energy recovery amplifier 800 ofFIG. 8 has been described with reference to two power amplifiers 820A,820B, such is not intended as limiting and one of ordinary skill in theart will readily appreciate that more than two power amplifiers may beused, and that one may choose to use additional resistance compressionnetworks 860 and rectifiers 865 to recover additional energy that wouldotherwise be dissipated in the power combining process. Moreover,although operation of the outphasing energy recovery amplifier 800 ofFIG. 8 has been shown with energy recovery directly to a power supplyapplied to the two amplifiers 820A, 820B, one of ordinary skill in theart will readily appreciate that energy may be recovered to any otherstorage location that may be convenient, and that power supplies for thepower amplifiers 820 could be derived elsewhere (e.g., for AMOmodulation).

Referring now to FIG. 9, in some embodiments an RCN 960 includes a firstRCN element 960A and a second RCN element 960B, characterized by aresistive input characteristic that varies little as the input powerchanges. The first RCN element 960A includes a first conjugate reactance962A in series with a first matched load resistance 964A and the secondRCN element 960B includes a second conjugate reactance 962B in serieswith a second matched load resistance 9648. First and second RCNelements 960A, 960B represent an equivalent resistance of two rectifiersas given by equation (6). The reactive branches are designed to have thespecified reactance X at the designed operating frequency. It can beshown that at this frequency the input impedance of the RCN 960 will beresistive with a value R_(RCN) indicated as follows:

$\begin{matrix}{R_{RCN} = {\frac{X^{2}}{2R_{rect}}\left\lbrack {1 + \left( \frac{R_{rect}}{X} \right)^{2}} \right\rbrack}} & (7)\end{matrix}$

In this way, compression of matched load resistances R_(rect) isprovided about a center value of impedance X. For variations of R_(rect)over a range having a geometric mean of X (i.e., R_(rect)ε[X/c_(rect)^(1/2)), c_(rect) ^(1/2)X], where c_(rect) is the ratio of the largestto smallest resistances in the R_(rect) range), the corresponding ratioof the compressed R_(RCN) range can be shown to be as follows:

$\begin{matrix}{c_{RCN} = \frac{1 + c_{rect}}{2\sqrt{c_{rect}}}} & (8)\end{matrix}$

For example, a 10:1 variation in R_(rect) (c_(rect)=10) results in amodest 1.74:1 variation in R_(RCN). Since R_(rect) is inverselyproportional to P_(in) as shown in equation (6), this means a 10:1variation in power delivered to the isolation port would result in onlya 1.74:1 variation in isolation port resistance. This narrowed range ofresistance will result in substantially improved isolation between theoutphased power amplifiers (as may be similar to outphased poweramplifiers 820A, 820B described in conjunction with FIG. 8), greatlyimproving amplification efficiency.

It should be noted that at sufficiently high output power levels (i.e.,low power levels to the rectifiers), the rectifier resistance can nolonger be effectively compressed. This is because at low input powerlevels, the diodes will be unable to turn “on” and overcome thecombination of supply voltage and diode built-in potential. When thediodes turn “off”, equations (5) and (6) are no longer valid and theefficiency of the RCN drops considerably. However, this poses no seriousproblems. In this region of operation, most of the power from the poweramplifiers is delivered to the load, and so the isolation port acts as avirtual open circuit. Therefore, the rectifier impedance and theefficiency of the RCN do not matter.

Referring now to FIG. 10, an RF transmission method 1000 includes, in apower supply, providing voltages to a switch network 1002, in the switchnetwork, outputting selected ones of the voltages to power amplifiers,two or more of the outputted voltages capable of being different 1004,and, in an RF combiner, combining power amplifier outputs and providingan RF circuit output signal 1006. In a further embodiment, the method1000 includes providing control voltages to the switch network 1010 andproviding phase-adjusted signals to RF input ports of the poweramplifiers 1012.

In a further embodiment, the method 1000 includes decreasing adifference between a sum of the powers outputted by the power amplifiersand an RF power outputted at the output port of the RF circuit. In stilla further embodiment, the method 1000 includes minimizing the differencebetween the sum of the powers outputted by the power amplifiers and theRF power outputted at the output port of the RF circuit.

In a further embodiment, the method 1000 includes gating on a variablenumber of transistors in at least one of the power amplifiers.

In a further embodiment, the method 1000 includes, in the RF combinercircuit, providing isolation between the plurality of input ports.

In a further embodiment, the method 1000 includes processing at least aportion of the RF power output from the power amplifiers using at leastone resistance compression network and at least one rectificationcircuit coupled to the at least one resistance compression network,wherein the processed RF power includes recovered RF power from the RFpower combiner circuit.

Having described embodiments of the concepts, circuits, and techniquesdescribed herein, it will now become apparent to one of ordinary skillin the art that other embodiments incorporating these concepts,circuits, and techniques may be used. It is felt therefore that theseembodiments should not be limited to disclosed embodiments, but rathershould be limited only by the spirit and scope of the appended claims.

What is claimed is:
 1. A radio-frequency circuit comprising: a pluralityof power amplifiers each having an input port, an output port and apower supply port; a power combiner having a plurality of input portsand an output port, said power combiner input ports coupled to theoutput ports of said power amplifiers; a means for providing a pluralityof voltages; and a means for selectively coupling ones of the pluralityof voltages to the power supply ports of said power amplifiers, whereinthe voltages provided to at least two of the power supply ports may beindependently selected from among the plurality of voltages.
 2. Theradio-frequency circuit of claim 1, wherein said means for selectivelycoupling ones of the plurality of voltages to the power supply ports ofsaid power amplifiers comprises a switching network.
 3. Theradio-frequency circuit of claim 2, wherein said means for selectivelycoupling ones of the plurality of voltages to the power supply ports ofsaid power amplifiers further comprises one or more low pass filterscoupled between the switching network and the power supply ports of theplurality of power amplifiers.
 4. The radio-frequency circuit of claim3, wherein said low-pass filters suppress undesired high-frequencycontent from being introduced at the output of said combiner.
 5. Theradio-frequency circuit of claim 2, wherein circuit parasitic inductanceand capacitance provide filtering to suppress undesired high-frequencycontent from being introduced at the output of said combiner.
 6. Theradio-frequency circuit of claim 1, wherein said means for providing aplurality of voltages comprises a power supply configured to provide aplurality of voltages from a voltage supply.
 7. The radio-frequencycircuit of claim 6, wherein said power supply is a switched-capacitorpower supply.
 8. The radio-frequency circuit of claim 6, wherein saidpower supply comprises: a power control circuit that produces a variablevoltage; and a switched-capacitor circuit coupled to the variablevoltage which produces said plurality of voltages.
 9. Theradio-frequency circuit of claim 1, further comprising: a controlcircuit configured to receive digital signals I and Q, and to producephase-modulated RF signals to the respective ones of the power amplifierinput ports.
 10. The radio-frequency circuit of claim 9, wherein saidcontrol circuit is further configured to produce signals to control saidmeans for selectively coupling ones of the plurality of voltages to thepower supply ports of said power amplifiers.
 11. The RF circuit of claim1, wherein: said voltage providing means comprises a power supply togenerate the plurality of voltages; and said means for coupling selectedones of the plurality of voltages corresponds to a switch network. 12.The RF circuit of claim 1, wherein each of the power amplifiers has anRF input port and further comprises means for controlling an outputsignal of the RF circuit by adjusting phases of the signals received atthe RF input ports and dynamically selecting the ones of the pluralityof voltages outputted to each of the power amplifiers.
 13. The RFcircuit of claim 1 further comprising means for decreasing a differencebetween a sum of the powers outputted by the plurality of RF amplifiersand an RF power outputted at the output port of the RF circuit.
 14. TheRF circuit of claim 1, wherein said means for combining comprises meansfor isolating each of the plurality of input ports.
 15. The RF circuitof claim 1, wherein said means for combining comprises: at least oneresistance compression network; and at least one rectification circuitcoupled to the at least one resistance compression network.
 16. Aradio-frequency (RF) circuit having an input port for receiving an RFsignal and having an output port at which an RF signal is provided, theRF circuit for coupling a plurality of voltages to a load and the RFcircuit comprising: a plurality of RF amplifiers each having an inputport coupled to the input of the RF circuit an output port and a biasterminal; a power combiner having a plurality of input ports and anoutput port with each of the plurality of power combiner input portscoupled to the output port of a corresponding one of said plurality ofRF amplifiers and with the power combiner output port coupled to theoutput port of the RF circuit; and a switch network for selectivelycoupling ones of the plurality of voltages to the bias terminals of saidRF amplifiers, wherein the voltages provided to at least two of the biasterminals are independently selected from among the plurality ofvoltages.
 17. The radio-frequency circuit of claim 16, wherein: said RFamplifiers are provided as RF power amplifiers; and the bias terminalsof said RF power amplifiers correspond to one or more power supply portsof said power amplifiers.
 18. The radio-frequency circuit of claim 17,wherein said switch network for selectively coupling ones of theplurality of voltages to the power supply ports of said power amplifiersfurther comprises one or more filters having a low pass frequencycharacteristic coupled between said switching network and the powersupply ports of said plurality of power amplifiers.
 19. Theradio-frequency circuit of claim 18, wherein said low-pass filters areprovided having filter characteristics selected to suppress undesiredhigh-frequency content from being introduced at the output of said powercombiner.
 20. The radio-frequency circuit of claim 17, wherein theradio-frequency circuit is provided having a circuit parasiticinductance and having a circuit parasitic capacitance which providesfiltering to suppress undesired high-frequency content from beingintroduced at the output of said power combiner.
 21. The radio-frequencycircuit of claim 16, further comprising: a control circuit configured toreceive digital signals I and Q, and to produce phase-modulated RFsignals to the respective ones of the power amplifier input ports. 22.The radio-frequency circuit of claim 21, wherein said control circuit isfurther configured to produce signals to control said means forselectively coupling ones of the plurality of voltages to the powersupply ports of said power amplifiers.
 23. The RF circuit of claim 16,wherein the switch network is configured to output selected ones of theplurality of voltages from the plurality of switch network output ports,at least two of the switch network output port voltages capable of beingdifferent ones of the plurality of voltages.
 24. The RF circuit of claim16 further comprising an RF combiner circuit having a plurality of inputports coupled to RF output ports of the plurality of RF amplifiers andan output port at which is provided an output signal of the RF circuit.25. The RF circuit of claim 18, wherein the power combiner circuitfurther comprises: a resistance compression network; and a rectificationcircuit coupled to the resistance compression network.
 26. The RFcircuit of claim 16 further comprising a control system configured toprovide the phase-adjusted signals over a plurality of first outputports coupled to the RF input ports of the plurality of RF amplifiersand the plurality of control signals over a plurality of second outputports coupled to the switch network.
 27. The RF circuit of claim 20,wherein the control system is further configured to decrease adifference between a total of the power output from the plurality ofpower amplifiers and a power output from the RF circuit.
 28. Aradio-frequency (RF) circuit comprising: a plurality of RF poweramplifiers each having an input port, an output port and a power supplyport; a power combiner having a plurality of input ports and an outputport, said power combiner input ports coupled to the output ports ofsaid power amplifiers; a switched-capacitor power supply for providing aplurality of voltages; and a switch network for selectively couplingones of the plurality of voltages between said switched-capacitor powersupply and the power supply ports of said RF amplifiers, wherein thevoltages provided to at least two of the power supply ports may beindependently selected from among the plurality of voltages.
 29. Theradio-frequency circuit of claim 28, wherein said switched-capacitorpower supply comprises: a power control circuit that produces a variablevoltage; and a switched-capacitor circuit coupled to the variablevoltage which produces said plurality of voltages.
 30. Theradio-frequency circuit of claim 29, further comprising: a controlcircuit configured to receive digital signals I and Q, and to producephase-modulated RF signals to the respective ones of the power amplifierinput ports.
 31. The radio-frequency circuit of claim 28, wherein saidcontrol circuit is coupled to said switch network and configured toproduce signals to control said switch network for selectively couplingones of the plurality of voltages to the power supply ports of saidpower amplifiers.
 32. A radio frequency (RF) circuit comprising: aplurality of RF amplifiers, each of the plurality of RF amplifiershaving an RF output and a supply voltage input; supply voltage means forproviding a plurality of amplifier supply voltages; means for couplingselected ones of the plurality of amplifier supply voltages provided bysaid supply voltage means to respective ones of the supply voltageinputs of the plurality of RF amplifiers, at least two of the amplifiersupply voltages capable of being different ones of the plurality ofamplifier supply voltages provided by said supply voltage means; andmeans for combining, having a plurality of input ports with each of theplurality of input ports coupled to a respective one of the RF outputports of said plurality of RF amplifiers, said means for combiningcapable of receiving RF signals from each of the plurality of RFamplifiers and combining the RF signals provided thereto to provide anRF output signal at an output thereof.
 33. A radio frequency (RF)circuit comprising: a power supply capable of generating a plurality ofvoltages; a switch network having a plurality of input ports coupled tothe power supply and a plurality of switch network output ports; aplurality of RF amplifiers, each having an RF output port and a supplyinput port coupled to a corresponding one of the plurality of switchnetwork output ports, wherein each of the RF amplifiers has an RF inputport configured to receive a phase adjusted signal and the switchnetwork is configured to receive at least one control signal, whereinthe phase adjusted signals and the at least one switching circuitcontrol signal are used to control the output signal of the RF circuit.